Transimpedance amplifier circuits, also referred to as current-voltage convertors or I-U convertors, are used in many areas—such as, for example, for photodetectors—for permitting precise measurements of small currents. Ideally, these amplifier circuits convert an input current into an output voltage proportional thereto. The ratio of output voltage Uout to input current Iin is designated as transimpedance ZEQ=Uout/Iin since the effect corresponds to an impedance.
A known possible design of a transimpedance amplifier is realized by one or more amplifier elements and a feedback branch. For example, operational amplifiers are used as amplifier elements, a customary basic operational amplifier circuit representing a transimpedance amplifier circuit. In the case of this basic circuit, the non-inverting input of an operational amplifier is connected to ground, and an input current signal is present at the inverting input of the operational amplifier. The operational amplifier is coupled with negative feedback to an ohmic resistance which determines the amplification factor. Ideally, the transimpedance ZEQ consists only of the ohmic resistance ZEQ=R and the amplification factor of the operational amplifier is infinite. In the case of the ideal transimpedance amplifier, it is remarkable that its function is frequency-independent. The signal current Iin at the input and the signal voltage Uout at the output could then have any desired frequency, so that, for example, information from a CD scanned via laser could be read out with very high frequencies and therefore very rapidly.
In the case of the real transimpedance amplifier, on the other hand, some non-ideally functioning factors have to be taken into account. For example, the ohmic feedback resistance has a parasitic self-capacitance which can be considered as a capacitance connected in parallel. This parallel capacitance limits the frequency bandwidth of the feedback resistance and hence that of the transimpedance amplifier.
Furthermore, photodiodes, avalanche photodiodes (APD) or CMOS sensors serving as a current source have a parasitic capacitance, in particular large-area APDs having a very high parasitic capacitance Cin. As a result of this capacitance Cin, no high bandwidths are permitted. The bandwidth BW is obtained as follows:
      BW    =          1              (                  2          ⁢          π          ⁢                                          ⁢                      C            in                    ⁢                                    R              EQ                        /            A                          )              ,in which REQ represents the transimpedance, A represents the linear amplification of the amplifier element and therefore REQ/A represents the input impedance of the transimpedance amplifier.
In addition, the real transimpedance amplifier also has a further, non-ideally functioning element—the amplifier element—such as, for example, the operational amplifier—itself. Thus, the real amplification, which is described by the ratio of the output to the input voltage, is not infinitely high, as is assumed in the case of the ideal transimpedance amplifier. Furthermore, the limited bandwidth of the amplifier element constitutes one of the main limits of a transimpedance amplifier.
The problem of the real transimpedance amplifier is therefore that its function is frequency-dependent. However, it would be desirable for it to operate in a frequency-independent manner in order to avoid distorting the information. However, this is possible in reality only in a limited frequency interval, a higher bandwidth which is antiproportionally dependent on the resistance value and on the parallel capacitance and in which the transimpedance amplifier operates in an approximately frequency-independent manner being realizable by smaller feedback resistances.
However, smaller feedback resistances result in a higher current noise Inoise, which depends on the resistance value as follows:
      I    noise    =                    4        ⁢        kT                    R        f            in which Rf represents the feedback resistance, T represents the absolute temperature and k represents the Boltzmann constant. The current noise due to the feedback resistance is disadvantageous since it is superposed on the input current which is generated, for example, by a photodiode and is to be actually converted by the transimpedance amplifier into a readily measurable output voltage and cannot differ therefrom.
It is true that a small current noise could now be achieved by the choice of a high feedback resistance. However, a high feedback resistance results—as mentioned above—in a relatively small bandwidth in which the transimpedance amplifier has the desired frequency-independent properties. Thus, it is desirable to make a so-called bandwidth-versus-noise compromise which is as optimal as possible.
For example, it is known that the feedback resistance can be replaced by a plurality of resistances which are connected in series. However, this technique is fairly quickly limited by the parallel capacitances to earth and the resulting length of the resistance chain.
The prior art also discloses the replacement of the feedback resistance by a feedback network, in particular a T-shaped network, which is composed, for example, exclusively of resistances. T-shaped feedback networks having exclusively capacitive components are also known, which is disclosed, for example, in the laid-open application WO 02/46779 A1. An overview of amplifier circuits of the prior art is also provided in “Photodiode Amplifiers” by Jerald G. Graeme, McGraw-Hill Publishers, USA 1996, pages 21 to 30.
The laid-open application U.S. Pat. No. 5,455,705 describes a transimpedance amplifier for an optical receiver having a photodetector generating a current and an integrating member for receiving the current, the integrating member having a capacitor. The output of the integrating member is connected to an amplifier stage which is formed for providing a voltage dependent on the output of the integrating member. A feedback resistance is connected between the output of the amplifier stage and the input of the integrating member and therefore determines both the nominal amplification and—in combination with the capacitor—the bandwidth of the transimpedance amplifier. For increasing the bandwidth, it is proposed to increase the amplification factor of the amplifier stage, in particular the amplification factor being made to be adjustable by a variable resistance in the amplifier stage. However, a current noise is not taken into account in this laid-open application.
Furthermore, it is known that transimpedance amplifiers having high transimpedance generate output signals with high amplitudes which may prove to be disadvantageous. In addition, high transimpedances require high open-circuit gains of the amplifier element.
A transimpedance amplifier circuit having a broad frequency band and a low noise and having a relatively low transimpedance REQ, in particular having a current noise of Inoise<root (4kT/REQ) dependent on the transimpedance, is therefore desirable.
Particularly in the case of transimpedance amplifier circuits which are used in photodetectors for laser rangefinders, non-distorting and low-noise conversion into a measurable output voltage is desirable for different and in particular relatively high frequencies of an input current generated by a photosensitive receiving element.